Radar system with improved angle formation

ABSTRACT

A radar system for recording the environment of a motor vehicle includes at least two transmitter antennas for emitting transmission signals, one or more receiver antennas for receiving transmission signals that have been reflected by objects, and signal processing equipment for processing the received signals. The antennas are arranged so that a phase center of at least one receiver antenna, with regard to a spatial direction R, does not lie outside of phase centers of two transmitter antennas that are offset in this spatial direction. The signals received by this receiver antenna are separated according to the two signal portions respectively originating from these two transmitter antennas.

FIELD OF THE INVENTION

The invention relates to a radar system for use in driver assistancesystems in the motor vehicle. According to invention the radar systemhas an improved angle formation, in particular in azimuth direction.

BACKGROUND INFORMATION State of the Art

Motor vehicles are increasingly equipped with driver assistance systems,which with the aid of sensor systems detect the environment and from thethus recognized traffic situation derive automatic reactions of thevehicle and/or instruct, especially warn the drivers. Here, adistinction is made between comfort and safety functions.

As a comfort function FSRA (Full Speed Range Adaptive Cruise Control)plays the most important roll in the current development. The vehicleadjusts the true speed to the desired speed predefined by the driver,provided the traffic conditions permit this, otherwise the true speed isautomatically adapted to the traffic situation.

In addition to an increase of the comfort, safety functions areincreasingly the focus, whereby the reduction of the or braking and/orstopping distance in emergency situations plays the most important role.The spectrum of the corresponding driver assistance functions extendsfrom an automatic priming of the brake to for reducing the brake latency(pre-fill), via an improved brake assistant (BAS+) up to the autonomousemergency braking.

For driver assistance systems of the above described type radar sensorare mainly used today. Also at poor weather conditions they workreliably and can measure in addition to the distance of objects alsodirectly their radial relative speed via the Doppler effect. Astransmission frequencies, here 24 and 77 GHz are used.

Since it is important for long-reach functions such as FSRA andcollision warning to form narrow antenna beams (to be able to determinethe lateral location of objects in sufficiently accurate manner and toseparate the objects well enough), there today preferably 77 GHz-sensorsare used—with 24 GHz the sensors would have to be about three times aswide with same azimuth beam focusing and unchanged antenna concept,which is very critical for the installation in the vehicle. On the otherside, however, the 77 GHz-technology today is still more expensive thanthe 24 GHz-technology.

SUMMARY OF THE INVENTION

It is, therefore, the object of the invention to achieve a high beamfocusing using a compact radar sensor in order to be able to realize inparticular long-reach functions in a vehicle.

This object has been achieved according to the invention in a radarsystem as set forth herein. The inventive radar system demonstrates howto nearly double the effective aperture of the sensor by providingplural transmitter antennas. Thus, as seen from the antenna, the sensorwidth is nearly twice as large as its real or actual width.

The advantages of the invention result from the fact that the sensor canbe reduced in physical size, which leads in particular to the fact thatalso for long-reach functions the favorable 24 GHz-technology can beused. Apart from this, a reduction of sensor size on the one handachieves a reduction of the sensor weight (which e.g. reduces the fuelconsumption of the vehicles) and on the other hand achieves a principalcost reduction (because of smaller dimensions of mechanical parts andcircuit boards, which has a very favorable effect with planar antennaswith the expensive high frequency substrates).

The radar system according to invention for detecting the environment ofa motor vehicle comprises transmission means for emitting transmissionsignals using at least two transmitter antennas, receiving means forreceiving transmission signals reflected by objects using one or morereceiver antennas, wherein each antenna comprises a phase center.Furthermore, signal processing means are provided for processing thereceived signals.

The phase center of at least one receiver antenna does not lie, withregard to a spatial direction R (e.g. R is horizontal), outside phasecenters of two transmitter antennas that are offset in said spatialdirection (e.g. for horizontal spatial direction not on the left or onthe right), and for one of these receiver antennas the signals receivedby said antennae are separated according to the two portions originatingfrom said two transmitter antennas. By means of this for this spatialdirection the effective antenna aperture can be increased and thus theaccuracy and resolution of the angular measurement performed in thesignal processing means can be improved.

Preferably, the antennas of the radar system arranged outside inrelation to the spatial direction R are used for transmission. In anadvantageous embodiment of the radar system the antennas are constructedin planar technology, e.g. as patch antennas.

In a further embodiment of the invention at least one antenna is usedfor both transmission and reception. The antenna works simultaneously astransmission and receiver antenna or its function changes temporallybetween transmission and receiver antennas.

In a positive embodiment of the invention the separation of the receivedsignals in portions originating from different transmitter antennas isrealized by temporal multiplex of these transmitter antennas, i.e. bycorresponding switching at any receiving time only one transmitterantenna each is active, wherein for objects moved relative to the radarsystem the resulting phase offset of received signals, which originatefrom different transmitter antennas, is considered for the angularmeasurement.

In particular, the separation of the received signals into portions,which originate from different, but simultaneously emitting transmitterantennas, is realized by the fact that for these different transmitterantennas at least one parameter is different for the modulation ofamplitude, frequency and/or phase of the emitted signal.

In particular, there are several transmitter antennas (number of N_(S),preferably N_(S)=2) and several receiver antennas (number of N_(E)),which each have at least approximately the same emission characteristicand which with regard to their phase centers in the spatial direction Rare each arranged at least approximately equidistantly, wherein in thisspatial direction R the distance of these N_(S) transmitter antennas toeach other is larger by an integral factor K≦N_(E) than the distance ofthese N_(E) receiver antennas to each other, whereby an arrangement witha transmitter antenna and maximum N_(S)·N_(E) receiver antennasequidistantly arranged in this spatial direction R is synthesized withan at least approximately identical emission characteristic.

In a further embodiment of the invention in the signal processing meansthe position of objects in the spatial direction R is determined by thefact that for the angle formation with regard to this spatial directiona digital beam formation (e.g. with a discrete Fourier transformation)or high-resolution methods (e.g. model-based methods such as accordingto Burg or subspace methods such as MUSIC) are used.

In a positive embodiment the N_(S) transmitter and the N_(E) receiverantennas are realized in planar technology and are arranged on a planesurface. Moreover, at least two of the N_(S) transmitter and of theN_(E) receiver antennas overlap with regard to the spatial direction R.This overlap is realized by at least one of the subsequent arrangementsand/or embodiments of these transmitter and receiver antennas:

-   -   Antennas are offset to each other with regard to the spatial        direction S which runs perpendicular to the spatial direction R,        e.g. for a horizontal spatial direction R the transmitter        antennas are arranged above the receiver antennas or vice versa.    -   The transmitter and/or the receiver antennas have an inclined        form, with regard to the spatial direction R, e.g. gaps and/or        the lines of the respective planar antennas are arranged not        parallel or vertical, but inclined, i.e. at an angle α with        0°|α|<90°, to the spatial direction R.    -   Antennas are locked with each other with regard to the spatial        direction R, as it is shown in a particular embodiment in FIG.        24.    -   Emitting and/or receiving elements (e.g. patches) are jointly        used by at least two transmitter and/or receiver antennas.    -   At least one antenna is used both for transmission and        reception.

In particular with regard to the spatial direction R the width of theN_(S) transmitter antennas and/or the width of the N_(E) receiverantennas is larger than the distance of the N_(E) receiver antennas toeach other (e.g. approx. by the factor 2), so that the emissioncharacteristic of these antennas is so narrow that the effect ofambiguities in the angle formation is reduced or as far as possibleavoided.

In an advantageous embodiment of the invention received signals areacquired from different combinations of the N_(S) transmitter and of theN_(E) receiver antennas, wherein the relative phase centers of thesecombinations of transmitter and receiver antennas lie at leastapproximately equidistant with regard to the spatial direction R. Here,the relative phase center of a combination of a transmitter and of areceiver antenna is defined as sum of the two vectors from a referencepoint to the phase centers of the transmitter and of the receiverantenna. In addition, with regard to the spatial direction S which isvertical to spatial direction R the position of the relative phasecenters of these combinations from transmitter and receiver antennasvaries periodically with the period length P, if a sequence of thesecombinations from transmitter and receiver antennas is considered, whichis arranged in the spatial direction R with regard to the position ofthe relative phase centers. An exemplary embodiment of such an antennaassembly is shown in FIG. 13 with the reference numeral 13.1.Furthermore, in the signal processing means the fact is utilized thatthe received signals of an object dependent from its angular position inthe spatial direction S have a phase portion alternating periodicallywith the period length P over the combinations of transmitter andreceiver antennas arranged in such manner, wherein for this spatialdirection S assertions on the angular position of objects and/or on themisalignment of the radar system are possible.

In a further embodiment of the radar system according to invention, inwhich the spatial direction R lies horizontal and the spatial directionS vertical a measure is used in the signal processing means fordetecting in particular stationary objects which can be passed fromabove or underneath. This measure utilizes at least one deviation withregard to the proportions resulting for received signals from only oneelevation angle. Wherein in connection with the reflecting property ofroad surfaces the size and/or the amount, in particular each filteredover the object distance, and/or the distance-related variation of thismeasure is used for an at least rough estimate of the height of objectsabove the road surface.

In an advantageous embodiment of the invention received signals areacquired from different combinations of transmitter and receiverantennas. The transmitter and receiver antennas used here each have atleast approximately the same emission characteristic, wherein theemission characteristic of these transmitter antennas can be differentfrom the emission characteristic of these receiver antennas. With regardto the spatial direction R the position of the relative phase centers ofthese combinations from transmitter and receiver antennas variesperiodically with the period length Q by an equidistant raster. In thesignal processing means for determining the position of objects in thespatial direction R the fact is utilized that the received signals of anobject, recorded by different combinations of transmitter and receiverantennas, dependent from its angular position in the spatial direction Rhave a phase portion alternating with the period length Q apart from alinear phase portion, if a sequence of the combinations from transmitterand receiver antennas ordered in the spatial direction R with regard tothe position of the relative phase centers is considered. The linearphase portion of the received signals allows for a fine, but ambiguousangle determination in the spatial direction R, whereas the alternatingphase portion allows for a rough, but clear angle determination.

BRIEF DESCRIPTION OF THE DRAWINGS

In FIG. 1 the first form of embodiment of a radar system is shown.

FIG. 2 shows for the first form of embodiment the frequency of thetransmission and of the received signals, which consist of so-calledfrequency ramps, as well as the antenna combinations sequentially driventhereby.

FIG. 3 shows a sampled signal with the presence of two objects beforethe first DFT (left) and after the first DFT (right).

In FIG. 4 the complex spectral value rotating via the frequency ramps inthe distance gate 4, in which there is exactly one object, is shown.

FIG. 5 shows the two-dimensional complex-valued spectrum after thesecond DFT.

FIG. 6 shows for the antenna assembly of the first form of embodimentthe different path lengths between the individual antennas and a faraway object resting relative to the sensor with an azimuth angleα_(Az)<0.

FIG. 7 a shows an antenna assembly with transmitter and 8 receiverantennas, which is to equivalent the antenna assembly of the first formof embodiment with 2 transmitter and 4 receiver antennas; in FIG. 7 bfor this equivalent arrangement the different path lengths between theindividual antennas and a far away object resting relative to the sensoris shown.

FIG. 8 a shows for the above antenna assemblies the complex spectralvalue rotating via the antenna combinations in thedistance-relative-speed-gate (9, 0), in which there is exactly oneobject (resting relative to the sensor); in FIG. 8 b the amount ofassigned spectrum after the third DFT is shown.

FIG. 9 shows the data before the three-dimensional DFT (left) and thethree-dimensional complex-valued spectrum thereafter (right).

In FIG. 10 the second form of embodiment of a radar system is shown.

FIG. 11 shows for the second form of embodiment the frequency of thetransmission and of the received signals with a parallel control of allantenna combinations.

FIG. 12 shows the third form of embodiment of a radar system.

FIG. 13 shows the fourth form of embodiment of a radar system.

FIG. 14 shows for the antenna assembly of the fourth form of embodimentthe different path lengths between the individual antennas and a faraway object resting relative to the sensor with an elevation angleα_(El)<0 and the azimuth angle α_(Az)<0.

FIG. 15 a shows for the proportions indicated in FIG. 14 the complexspectral value rotating over the antenna combinations in thedistance-relative-speed-gate (9, 0), in which there is exactly oneobject (resting relative to the sensor); in FIG. 15 b the amount of theassigned spectrum after the third DFT is shown with two power peaksdistanced by half the DFT length N=16 with n=2 and n=10.

FIG. 16 shows in the complex plane the connections for thetransformation of a relation for the elevation measuring capability ofthe fourth form of embodiment.

FIG. 17 illustrate the reflecting effect of the road surface.

FIG. 18 a shows the relative imaginary part of the proportion V and FIG.18 b its absolute imaginary part each along the distance r for a vehicle(average height h_(o)=0.5 m, vertical expansion 0.3 m) and a bridge(average height h_(o)=5 m, vertical expansion 0.3 m), wherein the sensorheight is h_(s)=0.5 m.

In FIG. 19 the connection Δφ=3π·sin(α_(Az)) resulting for d=3λ/2 betweenphase difference Δφ and azimuth angle α_(Az) is shown.

FIG. 20 a shows an ideal antenna diagram with azimuthal detection range−19.5 . . . +19.5; FIG. 20 b shows a realizable antenna diagram for sucha detection range with at least 15 dB suppression outside.

FIG. 21 shows the antenna assembly of a fifth form of embodiment of aradar system.

FIG. 22 shows the antenna assembly of a sixth form of embodiment of aradar system.

FIG. 23 shows the antenna assembly of a seventh form of embodiment of aradar system.

FIGS. 24 a and 24 b show two alternative antenna assemblies of an eighthform of embodiment of a radar system.

FIG. 25 shows the antenna assembly of a ninth form of embodiment of aradar system.

FIG. 26 shows the antenna assembly of a tenth form of embodiment of aradar system.

FIG. 27 shows the eleventh form of embodiment of a radar system.

FIG. 28 shows an antenna diagram, which is sensitive by correspondingshoulders also outside of the azimuthal range −19.5 . . . +19.5.

FIG. 29 shows for the eleventh form of embodiment a course of the amountof the spectrum after the third DFT with two power peaks distanced byhalf the DFT-length N=16 with n=2 and n=10.

DETAILED DESCRIPTION OF EXAMPLE EMBODIMENTS OF THE INVENTION

The invention is now explained on the basis of exemplary embodiments ofradar systems and arrangements of their antennas. First of all, itshould be noted that for images, diagrams and derivations point-shapedobjects are assumed, unless explicitly extended objects are pronounced.

Embodiment 1 According to FIG. 1

At first, the exemplary embodiment of a radar system, which is roughlyshown in FIG. 1 is considered. The radar system has 2 transmitterantennas TX0 and TX1 for emitting transmission signals and 4 receiverantennas RX0-RX3 for receiving transmission signals reflected atobjects; the antennas are embodied as patch antennas on a planar board1.1 in planar technology, wherein this board is oriented with regard tohorizontal and vertical direction as is shown in the drawing. Allantennas (transmitter and receiver antennas) have the same emissioncharacteristic in elevation and azimuth. The 4 receiver antennas (andthus their phase, i.e. emission centers) each have the same lateral,i.e. horizontal distance D=λ/2=6.2 mm to each other, wherein λ=c/24.15GHz=12.4 mm is the mean wavelength of the emitted signals; thehorizontal distance of the two transmitter antennas to each other is 4times as large, i.e. it amounts to 4 d=2λ.

Via the multiplexers 1.3 and 1.4 in each case one of the two transmitterantennas and one of the 4 receiver antennas can be selected.

The transmission signals emitted on the respectively selectedtransmitter antenna are gained from the high frequency oscillator 1.2 inthe 24 GHz-range, which can be changed in its frequency via a controlvoltage v_(control); the control voltage 1.9 is generated in the controlmeans. The signals received from the respectively selected receiverantenna are equally down-mixed in the real-valued mixer 1.5 with thesignal of the oscillator 1.2 into the low frequency range. Thereafter,the received signals go through a bandpass filter 1.6 with the showntransfer function, an amplifier 1.7 and an A/D converter 1.8;subsequently they are further processed in a digital signal processingunit 1.10.

So that the distance of objects can be measured,—as is shown in FIG.2—the frequency of the high frequency oscillator and thus of thetransmission signals is changed very quickly in linear manner (in 8 μsby 187.5 MHz); this is referred to as a frequency ramp. The frequencyramps are periodically repeated (all 10 μs); in total there are 2048frequency ramps. Via the frequency ramps the 8 combinations of the 2transmitter and 4 receiver antennas are periodically repeated in theorder TX0/RX0, TX0/RX1, TX0/RX2, TX0/RX3, TX1/RX0, TX1/RX1, TX1/RX2 andTX1/RX3, wherein before each frequency ramp the respective nextcombination is selected. In FIG. 2, k is the indexed variable over the2048/8=256 frequency ramps for each antenna combination andm=4·m_(TX)+m_(RX) the indexed variable over the 8 antenna combinationsTXm_(TX)/RXm_(RX).

The received signal of a single object is a sinusoidal oscillation aftermixture and thus also at the A/D converter for each frequency ramp andeach of the 8 antenna combinations; this can be explained with the aidof FIG. 2 as follows: if the object has the radial relative speed zeroto the radar system, then the frequency difference Δf between thetransmitted signal and the received signal is constant and proportionalto signal propagation time Δt and thus proportional to the radialdistance r=c Δt/2, wherein c is the speed of light and the factor ½considers that the propagation time Δt refers to the forth and backpropagation of the wave; the frequency difference Δf results with theabove interpretation to Δf=2r/c·187.5 MHz/8 μs=r·156.250 kHz/m. Sincethe received signal is real-valued mixed with the oscillator and thuswith the transmission frequency, a sinusoidal oscillation with thefrequency Δf results after the mixer. This frequency lies in theMHz-range and is still shifted with a non-vanishing radial relativespeed by the Doppler frequency, which lies, however, only in thekHz-range and which is, therefore, approximately negligible compared tothe frequency portion by the object distance. If there are severalobjects, then the received signal is a superposition of severalsinusoidal oscillations of different frequency.

During each frequency ramp the received signal is sampled in each case256 times at the A/D converter at the distance of 25 ns (i.e. with 40MHz) (see FIG. 2). As is apparent from FIG. 2, a signal sampling makessense only in the time range, where received signals of objects arrivewithin the receivable distance range—thus after the ramp start at leastthe propagation time corresponding to the maximum receivable distancemust be waited for (with a maximum receivable distance of 150 m thiscorresponds to 1 μs).

Then via the 256 sampled values of each frequency ramp a DiscreteFourier Transformation (DFT) in form of a Fast Fourier Transformation(FFT) is formed. This makes it possible to separate objects in differentdistances, which lead to different frequencies (see FIG. 3; on the leftsignal before DFT with presence of two objects, on the right after DFT).Each of the discrete frequency supporting points j of the DFTcorresponds to a distance r and can, therefore, analogue to pulse radarsalso be called a distance gate; with the above interpretation thedistance gates have merely one distance and thus a width of one meter(results from r·156.250 kHz/m=1/(6.4 μs)). In the distance gates, inwhich the objects are located, power peaks occur in the DFT. As thesampled received signals are real-valued and the upper transition regionof the analogue bandpass filter 1.5 in FIG. 1 has a frequency bandwidthof 8.75 MHz (corresponds to the range of 56 frequency supportingpoints), only 100 of the 256 discrete frequency supporting points can befurther processed (it should be noted that it is not possible to realizeany number of narrow transition regions of filters). The filter 1.5absorbs small frequencies and thus the received signals of closeobjects, in order to avoid an overmodulation of the amplifier 1.6 and ofthe A/D converter 1.7 (the signals received at the antennas get strongerwith decreasing object distance).

Over the 256 frequency ramps (k=0, 1, . . . , 255) complex spectralvalues e(j, k, m) result in each of the 8 antenna combinations m (m=0,1, . . . , 7) for each distance gate j (thus each of the 100 consideredfrequency supporting points). If there is exactly one object in thedistance corresponding to a distance gate, the complex spectral valuerotates in this distance gate j over the 256 frequency ramps of eachantenna combination with the Doppler frequency, since from frequencyramp to frequency ramp the distance (in the mm range or below) and thusthe phase position of the assigned oscillation change uniformly (seeFIG. 4; the phase change of 45° per frequency ramp represented therecorresponds to a distance decrease of the object of λ/(8−2)=0.78 mm,wherein the mean wavelength λ=c/24.15 GHz=12.4 mm and the factor 2 inthe denominator considers the forth and back propagation of the wave,resulting in the radial relative speed v_(rel)=0.78 mm/80 μs=35 km/h;positive sign of the radial relative speed is defined as anapproximation). Several objects with different radial relative speed inthe same distance gate are separated by the fact that a second DFT iscalculated for each antenna combination and each distance gate over thecomplex spectral values resulting in the 256 frequency ramps. Eachdiscrete frequency supporting point I of this second DFT corresponds toa set of Doppler frequencies (because of the sampling of the Dopplerfrequency it can be defined only up to an unknown integral multiple ofits sampling frequency) and thus to a set of radial relative speedsv_(rel) of objects, so that the discrete frequency supporting points ofthe second DFT can be referred to as relative-speed-gates; forlinguistic simplification from this point on the addition “radial” isomitted for the radial relative speed. The second DFT does not onlyserve for determining the relative speed, but it increases by itsintegration also the detection sensitivity—with 256 frequency rampsapproximately by 10⊙log₁₀(256)=24 dB.

After this second DFT for the relative speeds for each antennacombination a two-dimensional complex-valued spectrum results, whereinthe individual cells can be referred to as distance-relative-speed-gatesand wherein power peaks occur at the respectively assigneddistance-relative-speed-gate by objects (see FIG. 5).

Finally, then the information from the 8 antenna combinations is merged.The waves reflected at a single object and originating from the twotransmitter antennas arrive at the 4 receiver antennas dependent on theazimuth angle α_(Az) with different phase positions to each other, sincethe distances between object and transmitter and receiver antennas areslightly different. This is now explained in detail, wherein theconsidered object first is to rest relative to the sensor, i.e. it hasthe relative speed zero. In FIG. 6 in vertical projection the phasecenters of the antennas as well as the beam paths to a far away objectresting relative to the sensor are shown with an azimuth angle α_(Az)<0(positive α_(Az) means to the right of the perpendicular surface to theboard plane) and an elevation angle α_(El)=0 (in the horizontalperpendicular surface to the board plane); the object is so far awaythat the beam paths can be assumed to be parallel, this means the objectis in the far field of the antenna assembly. The path length r(m) forthe antenna combination m=4·m_(TX)+m_(RX) from the transmitter antennaTXm_(TX) to the object and back to the receiver antenna RXm_(RX) resultsinr(m)=2·r _(RP)+sin(−α_(Az))·(a+m _(TX)·4d+a+d/2+m _(RX) ·d)=2·r_(RP)+sin(−α_(Az))·(2a+d/2+m·d),wherein r_(RP) is the path length from a reference point RP on theantenna plate to the object and a is the horizontal distance between thereference point and the transmitter antenna TX0. From this relation itcan be seen that the distance changes linear with the number m of theantenna combination. The size (2a+d/2+m·d) represents the horizontaldistance of the so-called relative phase center of the antennacombination m to the reference point RP and is the sum of horizontaldistance of the assigned transmitter and receiver antenna to thereference point (the relative phase center of a combination of atransmitter and of a receiver antenna here is defined as a sum of thetwo vectors from a reference point to the phase centers of thetransmitter and of the receiver antenna).

The phase difference φ(m)−φ(0) between the receiving waves for theantenna combination m=0, 1, . . . , 7 and the antenna combination m=0results based on the different path lengths r(m) to

$\begin{matrix}{{{\varphi(m)} - {\varphi(0)}} = {{- 2}{{\pi/\lambda} \cdot \left\lbrack {{r(m)} - {r(0)}} \right\rbrack}}} \\{= {{- 2}{{\pi/\lambda} \cdot \left\lbrack {{2 \cdot r_{RP}} + {{\sin\left( {- \alpha_{Az}} \right)} \cdot \left( {{2a} + {d/2} + {m \cdot d}} \right)} -} \right.}}} \\\left. {{2 \cdot r_{RP}} - {{\sin\left( {- \alpha_{Az}} \right)} \cdot \left( {{2a} + {d/2} + {0 \cdot d}} \right)}} \right\rbrack \\{= {{{- 2}{{\pi/\lambda} \cdot {\sin\left( {- \alpha_{Az}} \right)} \cdot d \cdot m}} = {2{{\pi/\lambda} \cdot {\sin\left( \alpha_{Az} \right)} \cdot d \cdot m}}}}\end{matrix}$and thus changes likewise linear with the number m of the antennacombination. The amplitude of the signals received on the differentantenna combinations is constant, since all antennas have the sameemission characteristic and the distance from the antennas to the faraway object for a level consideration differs only slightly.

As it is directly obvious, for the antenna assembly represented in FIG.7 a with vertical projection according to FIG. 7 b exactly the samerelations for the path length r(m) and the phase difference φ(m)−φ(0)result as for the arrangement considered so far according to FIG. 1; thearrangement according to FIG. 7 a has only one transmitter antenna TX0and 8 equidistant receiver antennas RX0-RX7, wherein the antennacombination m=m_(RX) is now formed from the transmitter antenna and thereceiver antenna RXm_(RX). Based on identical individual antennas andidentical phase relations of the antenna combinations to each other bothantenna assemblies are equivalent with regard to the angle measuringcapability. However, the arrangement according to FIG. 1 representedhere has the advantage that it has nearly only half the horizontalexpansion compared to the conventional arrangement according to FIG. 7a, as a result of which the sensor size can be significantly reduced.

The azimuth angle-dependent phase differences φ(m)−φ(0) which are linearto and/or decreasing via the 8 antenna combinations m are remained untilthe second DFT apart from possible constant and thus phase shifts whichcan be compensated; this means that if there is only one object in adistance-relative-speed-gate (j, l), the respective complex spectralvalue (j, l, m) there rotates via the 8 antenna combinations m=0, 1, . .. , 7 with constant rotational speed dependent on the azimuth angle (seeFIG. 8 a as an example). Hence in each distance-relative-speed-gate adigital beam formation for the azimuth direction can be performed. Forthis purpose sums are created via the complex values to the 8 antennacombinations, which are each multiplied with a set of complex factorswith a linear changing phase; dependent on the linear phase change ofthe respective factor set lobes with different beam directions result.The beam width of these lobes is significantly smaller than that of theindividual antennas. The above described summation is realized by a16-point-DFT, wherein the 8 values of the 8 antenna combinations aresupplemented by 8 zeros. The discrete frequency values n=0, 1, . . . ,15 of this DFT correspond to different phase differencesΔφ=φ(m)−φ(m−1)=2π·n/16 between adjacent antenna combinations and thus todifferent azimuth angles α_(Az)=arcsin(Δφ·λ/(2 πd))=arcsin(n·λ/(16 d))and therefore can be referred to as angle gates. In FIG. 8 b the amountof the course w(j, l, n) of the spectrum of the third DFT is shown forthe proportions according to FIG. 8 a, which refer to an object belowthe azimuth angle α_(Az)=14.5° (n=2 corresponds to the represented phasedifference between adjacent antenna combinations of 45°, whichcorresponds to π/4, and for d=λ/2 the azimuth angleα_(Az)=arcsin(π/4)=14.5° corresponds). The third DFT does not only servefor determining the azimuth angle, but it increases by its integrationalso the detection sensitivity—with 8 antenna combinations approximatelyby 10·log₁₀(8)=9 dB.

So far it has been assumed for the determination of the azimuth anglethat the object has the relative speed zero. If this is not the case,the phase between the antenna combinations still changes additionally inlinear mode proportional to the relative speed, as the received signalsof the 8 successive antenna combinations in accordance with FIG. 2 havea respective time delay of 10 μs and the distance each changes slightlyduring this period. Since each third DFT belongs to adistance-relative-speed-gate and thus to a certain relative speed, thelinear phase change generated by the relative speed can be compensatedover the 8 antenna combinations either before or after the third DFT.With a compensation before the DFT the phase of the complex input valuesmust be shifted, with a compensation after the DFT the discretefrequency values n belonging to the output values must be shifted. Dueto the above explained ambiguities for the relative speed thiscompensation leads to different azimuth angles dependent on the usedhypothesis for the ambiguous relative speed.

After this third DFT for the azimuth angles (inclusive the compensationof the linear phase change generated by the relative speed over theantenna combinations) a three-dimensional complex-value spectrumresults, wherein the individual cells can be referred to asdistance-relative-speed-angle gates and power peaks can occur by objectsat the respectively assigned distance-relative-speed-angle-gate (seeFIG. 9; on the left data before three-dimensional DFT, on the rightthereafter). Thus, by determining the power peaks objects can bedetected and their measures distance, relative speed (apart frompossible ambiguities, see above) and azimuth angle (to each ambiguityhypothesis of the relative speed a value corresponds, see FIG. 9) can bedetermined. Since power peaks caused by the DFT-windowing still havelevels also in adjacent cells, the object dimensions can be determinedby interpolation in response of these levels still substantial moreaccurately than the gate width. It should be noted that the windowfunctions of the three DFTs are selected such that on the one hand thepower peaks do not get too wide (for a sufficient object separation),but on the other hand also the side lobes of the window spectra do notget too high (in order to be able to detect also weakly-reflectiveobjects in the presence of highly-reflective objects). From the heightof the power peaks as the fourth object dimension also its reflectioncross section can be estimated, which indicates, how strong the objectreflects the radar waves.

The described detection of objects and the determination of the assignedobject dimensions represent a measuring cycle and supply a momentaneouspicture of the environment; this is periodically repeated approx. all 30ms. For judging the environment situation the momentaneous pictures arepursued, filtered and evaluated throughout successive cycles; thereasons for this are in particular:

-   -   some sizes cannot be determined directly in a cycle, but only        from the change over successive cycles (e.g. longitudinal        acceleration and lateral speed),    -   the movement of objects can be checked for plausibility over        several cycles, resulting in a more robust and safer description        of the environment; thus e.g. the change of the distance        resulting over successive cycles must comply with the measured        (radial) relative speed, which results in redundancy and thus        additional safety in the description of the environment,    -   reduction of measuring noise by temporal filtration over several        cycles.

Pursuing and filtering of object detections over successive cycles isalso referred to as tracking. Here, for each object values for the nextcycle are predicted from the tracked object dimensions of the currentcycle. These predictions are compared with the objects and their objectdimensions detected in the next cycle as a snapshot, in order tosuitably assign them to each other. Then the predicted and measuredobject dimensions belonging to the same object are merged, from whichresult the current tracked object dimensions, which thus representvalues filtered over successive cycles. If certain object dimensionscannot be clearly determined in a cycle, the different hypotheses are tobe considered with the tracking. From the tracked objects and theassigned tracked object dimensions the environment situation for therespective driver assistance function is analyzed and interpreted, inorder to derive from it that or the relevant objects and thus thecorresponding actions.

Embodiment 2 According to FIG. 10

The embodiment of the sensor according to FIG. 1 considered so far hasthe disadvantage that the 8 antenna combinations are sequentiallyoperated, i.e. reception is always performed only on one antennacombination—this affects negatively the system sensitivity. Thearrangement according to FIG. 10 overcomes this disadvantage. Bothtransmitter antennas TX0 and TX1 are parallel operated, and the signalsof the 4 receiver antennas RX0-RX3 are parallel evaluated. For thispurpose, the output signal of the high frequency oscillator 10.2 isapplied over the power divider 10.3 simultaneously to both transmitterantennas, and on the receiving side up to the digital signal processingmeans there are 4 parallel channels. With each frequency ramp thus bothtransmitter antennas are used and the signals of all 4 receiver antennasare evaluated, wherein the frequency ramps and the sampling during thefrequency ramps are now temporally stretched by the factor 4, i.e. thenow 512 frequency ramps have a respective duration of 32 μs and areperiodically repeated all 40 μs, and the sampling of the 256 valueshappens with 10 MHz, i.e. all 100 ns (see FIG. 11).

In order to be able to separate in the received signals the portions ofboth transmitter antennas, before the transmitter antenna TX1 theswitchable inverters 10.4 is located (the switchable inverter isactivated from the control means 10.9). The switchable inverter isactive each second frequency ramp, i.e. with each second frequency rampthe phase position of the transmitter antenna TX1 is shifted by 180° inrelation to the other frequency ramps. With this the phase of thereceived signals, which are generated by transmission signals of TX1reflected at an object, alternates from frequency ramp to frequency rampby 180° in addition to the change by the relative speed of the object.Thus before the second DFT these received signals originating from TX1have an additional phase modulation of 180° with the period length 2,which leads with the second DFT to a displacement of the spectrum byhalf the DFT-length and thus 12.5 kHz.

The second DFT has now the length 512 (there are 512 frequency ramps)and is determined for the four reception channels and for each distancegate. By the phase modulation of TX1 an object in the second DFTgenerates in each reception channel and in the corresponding distancegate two power peaks at the distance of 12.5 kHz; the power peak at thefrequency corresponding to the relative speed originates from thetransmitter antenna TX0, the power peak shifted by 12.5 kHz originatesfrom the transmitter antenna TX1. Thus the portions originating from thetwo transmitter antennas are separated.

For the third DFT the 8 antenna combinations are generated by the factthat for each of the 4 reception channels (of the 4 receiver antennas)the lower half (0-12.5 kHz) of the second DFT for TX0 and the upper half(12.5 kHz-25 kHz) for TX1 is used, wherein the upper half is shifteddownward by 12.5 kHz on the same Doppler frequency range 0-12.5 kHz asthe lower half. Thus there are again only 256 relative speed gates aswith the original embodiment 1. After the third DFT it results as asingle difference to the original arrangement that different hypothesesfor the relative speed do no longer mean different azimuth angles, butalways the same azimuth angle results (in the data cuboid of thethree-dimensional DFT according to FIG. 9 the only change is that withthe azimuth angle always the values belonging to the relative speedrange 0-280 km/h are valid); this is reasoned by the fact that there isno more time delay between the received signals of the 8 antennacombinations.

Instead of the above explained alternate phase variation of TX1 by 180°it could be also arranged at random, i.e. from frequency ramp tofrequency ramp the state of the switchable inverter is selected atrandom. Then the second DFT would have to be determined twice, once withand once without correction of the phase variation. In the DFTcalculated with phase correction the received signals which originatefrom the transmitter antenna TX1 would lead to power peaks, whereas thereceived signals which originate from the transmitter antenna TX0 wouldproduce for instance a noise lying approx. 27 dB below them; in the DFTcalculated without phase correction the proportions would be exchanged.Thus again also a separation of both portions would be possible. Theunambiguous range of the relative speed would double here.

By using 4 parallel reception channels the system sensitivity increasesby 6 dB, since the bandwidth of the bandpass filters 10.6 in relation tothe original embodiment is reduced by the factor 4 (the sampling duringthe frequency ramps is slower by a factor 4, since the frequency rampsare longer by this factor). Based on the double length 512 of the secondDFT in addition this results in an integration gain higher by 3 dB. Forthe case that per transmitter antenna the same high power is emittedduring the frequency ramps, this results in a total increase of thesystem sensitivity by 9 dB. If the transmission power is halved pertransmitter antenna (e.g. due to the simultaneous supply of two antennasfrom one source or because the entire transmission line is limited dueto approval regulations), this results in an increase of the systemsensitivity by 6 dB.

Finally it should be mentioned that in the embodiment 2 according toFIG. 10 considered here the 4 parallel A/D converters could be replacedalso by a single A/D converter with an upstream multiplexer; this A/Dconverter would then work with the same cycle of 40 MHz as in theoriginal embodiment 1 according to FIG. 1.

Arrangement 3 According to FIG. 12

With the arrangements according to FIG. 1 and FIG. 10 considered so far,the transmitter antennas were arranged above the receiver antennas,since with an arrangement in one level (i.e. if the transmitter antennaswould have been moved downward) the patches would have contacted eachother, which cannot be realized. By the arrangement of the transmitterand receiver antennas one above the other, the latter can be onlyapprox. half as high with a predetermined sensor height as with anarrangement in a plane, which in elevation direction leads to a smallerbeam focusing and thus to a smaller antenna gain. By means of this onthe one hand the vertical detection range increases, which can bedisadvantageous for certain functions (e.g. because bridges can behardly distinguished from stationary vehicles), and to the other handthe system sensitivity is reduced.

This limitation in the arrangement of the antennas can be avoided by thefact that at least one antenna is used both for transmission and forreception. FIG. 12 shows as an example an arrangement, in which theright antenna is used as a transmitter antenna TX1 and receiver antennaRX3; the horizontal distance of the receiver antennas to each other isstill d=λ/2, that of the transmitter antennas is still 4 d=2λ.

If the right antenna cannot work simultaneously as a transmitter andreceiver antenna, but can only temporally change its function betweentransmitter and receiver antenna, then the eighth antenna combination oftransmitter antenna TX1 and receiver antenna RX3 is not possible, sothat there are only 7 antenna combinations; for the third DFT then thesignal of the eighth antenna combination is to be set to zero. In theother case, thus if the right antenna can work simultaneously as atransmitter and receiver antenna, all 8 antenna combinations arepossible.

So that an antenna can be used both for transmission and reception, thisantenna must be connected alternately or permanently with theHF-generation and the receiving mixer; this can be realized for examplewith the subsequent forms of embodiment for the connecting element12.11:

-   -   Multiplexer: antenna can change its function temporally between        transmitter and receiver antenna, but cannot transmit and        receive simultaneously; in the multiplexer this results        typically in power losses within the range of 3 dB,    -   Circulator: allows for simultaneous transmission and reception        without power losses, however is very expensive,    -   Coupler structures (e.g. ring coupler or Wilkinson divider):        equally allows for simultaneous transmission and reception,        however with power losses (3-4 dB), however, their costs are        negligible, since they consist only of printed structures.

If it comes to power losses in the used connecting element and they arenot compensated by a corresponding different transmission power, thereceiving signals of the different antenna combinations do not have thesame level; this is to be considered with the angle formation methods(e.g. with the digital beam formation) and is to be compensated, ifnecessary.

If in the above example according to FIG. 12 the right and the leftantenna could be used each for simultaneous transmission and reception,then 9 antenna combinations with horizontally equidistant relative phasecenters could be generated.

So that all antennas in their environment on the board see as similarproportions as possible, on the left and/or on the right side of theoutside antennas TX0 and TX1/RX3 in the distance d=λ/2 so-called blindantennas could be arranged with the same structure as the effectiveantennas (i.e. an antenna column with 8 patches); these blind antennaswould then have to be locked with adaptation. By way of this allantennas would be affected in same way by the respective neighborantennas (in particular by coupling), which is more uncritical for theangle formation methods than a different impact by neighbor antennas.

Embodiment 4 According to FIG. 13

For all the embodiments considered so far it was only possible tomeasure the azimuth angle of objects, however, not to measure theelevation angle. However, the latter would be favorable for manyfunctions, in particular if it is to be reacted to stationary objects onthe roadway (vehicles, pedestrians), in order to distinguish them fromobjects above the roadway (bridges, signs) or from small objects locatedon the roadway (e.g. can of coke) as well as from reflections of theroad surface (e.g. by uneven board joints).

In order to be able to measure and/or estimate the elevation angle, therelative phase centers of the antenna combinations must have a differentvertical position (the relative phase center of a combination of atransmitter and of a receiver antenna is here defined as sum of the twovectors from a reference point to the phase centers of the transmitterand of the receiver antenna). In the antenna assembly according to FIG.13 now considered as an example the two receiver antennas RX1 and RX3are offset downward by s=λ/2 in relation to the other two receiverantennas RX0 and RX2; otherwise the embodiment according to FIG. 13 doesnot differ from the original embodiment according to FIG. 1.

In FIG. 14 in horizontal projection (i.e. board 13.1 seen from the side)the phase centers of the antennas as well as the beam paths to a faraway object resting relative to the sensor with an azimuth angleα_(Az)=0 and an elevation angle α_(El)>0 (positive α_(El) means above)are shown (the object is so far away that the beam paths can be assumedto be parallel, i.e. the object is in the far field of the antennaassembly). The path length r(m) for the antenna combinationm=4·m_(TX)+m_(RX) from the transmitter antenna TXm_(TX) to the objectand back to the receiver antenna RXm_(RX) results in

$\begin{matrix}{{r(m)} = {{2 \cdot r_{RP}} + {{\sin\left( \alpha_{El} \right)} \cdot \left( {b + b + c + {{{mod}\left( {m_{RX},2} \right)} \cdot s}} \right)}}} \\{{= {{2 \cdot r_{RP}} + {{\sin\left( \alpha_{El} \right)} \cdot \left( {{2b} + c + {{mod}\;{\left( {m,2} \right) \cdot s}}} \right)}}},}\end{matrix}$wherein r_(RP) is the path length from a reference point RP on theantenna plate to the object, b is the vertical distance between thereference point and the transmitter antennas, c is the vertical offsetbetween transmitter antennas and the two upper receiver antennas RX0 andRX2 and mod(., 2) the modulo function to 2. The size (2b+c+mod(m, 2) s)represents the vertical distance of the relative phase center of theantenna combination m to the reference point RP and is the sum from thevertical distance of the assigned transmitter and receiver antenna tothe reference point.

The phase difference Δφ_(El)=φ(1)−φ(0) between the received waves andthus the received signals for the antenna combinations m=1, 3, 5, 7 tothe lower receiver antennas and for the antenna combinations m=0, 2, 4,6 to the upper receiver antennas results due to the different pathlengths r(m) toΔφ_(El)=−2π/λ·[r(1)−r(0)]=−2π/λ·sin(α_(El))·s.

The phase φ(m) of the received signals thus alternates with the periodlength 2 via the number m=0, 1, . . . , 7 of the antenna combinations bythis value αφ_(El). If now in addition a generally nonvanishing azimuthangle α_(Az) is considered, then the phase φ(m) of the received signalsadditionally comprises a portion linearly changing over the antennacombinations m (see in front) and results in total toφ(m)=φ(0)+2π/λ·sin(α_(Az))·d·m−2π/λ·sin(α_(El))·s·mod(m,2).

As far as in the assigned distance-relative-speed-gate (j, l) there isonly this one object, then the complex values v(j, l, m) there after thesecond DFT over the 8 antenna combinations m=0, 1, . . . , 7 result in

$\begin{matrix}{{v\left( {j,l,m} \right)} = {K \cdot {\exp\left\lbrack {j \cdot \left( {{\varphi(0)} + {2{{\pi/\lambda} \cdot {\sin\left( \alpha_{Az} \right)} \cdot d \cdot m}} -} \right.} \right.}}} \\\left. \left. {2{{\pi/\lambda} \cdot {\sin\left( \alpha_{El} \right)} \cdot s \cdot {{mod}\left( {m,2} \right)}}} \right) \right\rbrack \\{= {K \cdot {\exp\left\lbrack {j \cdot \left( {\varphi(0)} \right\rbrack \cdot {\exp\left\lbrack {{j \cdot 2}{{\pi/\lambda} \cdot {\sin\left( \alpha_{Az} \right)} \cdot d \cdot m}} \right\rbrack} \cdot} \right.}}} \\{{\exp\left\lbrack {{{- j} \cdot 2}{{\pi/\lambda} \cdot {\sin\left( \alpha_{El} \right)} \cdot s \cdot {mod}}\;\left( {m,2} \right)} \right\rbrack},}\end{matrix}$wherein K is the constant amount of these values and exp is theexponential function; the example according to FIG. 15 a resulting for asmall negative elevation angle α_(El) shows that compared with thecourse v(9, 0, m) represented in FIG. 8 a for the original embodiment 1now each second pointer is shifted by the phase Δφ_(El).

The additional factor f(m)=exp[−j·2π/λ·sin(α_(El))·s·mod(m, 2)],effected by the offset of the receiver antennas, changes the spectrumw(j, l, n) of v(j, l, m) formed in the third DFT as is explainedhereinafter. For even m=0, 2, . . . this factor has the value 1, for oddm=1, 3, . . . it has the value exp[−j·2π/λ·sin(α_(El))·s]. The spectrumF(n) of this factor alternating with the period length 2 has two powerpeaks (as one can easily derive from the transformation equation), apower peak with the frequency n=0 with the average of the two valuesmultiplied with the DFT-length N:F(0)=N/2·(1+exp[−j·2π/λ·sin(α_(El))·s]),and a second power peak with half the DFT-length, thus the frequencyn=N/2 with the difference of the two values multiplied with half theDFT-length:F(N/2)=N/2·(1−exp[−j·2π/λ·sin(α_(El))·s]).

The entire spectrum w(j, l, n) ensues by convolution of F(n) with theoriginal spectrum, which has a power peak with the frequency n_(Az)corresponding to the azimuth angle α_(Az); consequently it has two powerpeaks with the original frequency n_(Az) and the frequency n_(Az)+N/2,distanced by half the DFT-length, wherein for the proportion of thecomplex spectral values of these two power peaks it applies:w(j,l,n _(Az))/w(j,l,n _(Az)+N/2)=(1+exp[−j·2π/λ·sin(α_(El))·s])/(1−exp[−j·2π/λ·sin(α_(El))·s]).

FIG. 15 b shows as an example the amount of the spectrum w(9, 0, n) tothe course v(9, 0, m) represented in FIG. 15 a; apart from the powerpeak with n=n_(Az)=2 to the azimuth angle α_(Az)=14.5° there is a secondpower peak with n=10, i.e. half the DFT-length N=16 away.

From the proportion w(j, l, n_(Az))/w(j, l, n_(Az)+N/2) the elevationangle α_(El) can be determined. For this purpose the right side of theabove relation is described with the aid of FIG. 16 as follows:w(j,l,n _(Az))/w(j,l,n _(Az)+N/2)=exp(j·π/2)·cos(Δφ_(El)/2)/sin(Δφ_(El)/2)=j/tan(Δφ_(El)/2) withΔφ_(El)=−2π/λ·sin(α_(El))·s;by resolution of the elevation angle α_(El) it results for Δφ_(El) ε]−π, π[: α_(El)=arcsin(−λ/(π·s)·arctan [j·w(j, l, n_(Az)+N/2)/w(j, l,n_(Az))]).

Thus the elevation angle α_(El) in the range ]−arcsin(λ/(2 s),+arcsin(λ/(2 s)[ corresponding to Δφ_(El) can be clearly determined (forthe case s=λ/2 considered in the example it is thus in the range ]−90°,+90°[). However, this applies only under the condition that thefrequency value n_(Az) to the azimuth angle α_(Az) is known. Since,however, to an object there are generally two power peaks with half DFTlength distance, there are two hypotheses for the azimuth angle withrespectively different elevation angle. With the aid of the tracking(i.e. the observation of objects over several cycles) with the movementof the own vehicle it can be generally recognized, which hypothesis isthe right one, since only for a hypothesis a useful course of the objectplace results.

For ideal conditions (no noise and a point-shaped reflective object) inthe above function the argument of the arctan function is real-valued,for other conditions it generally has, however, still anothercomplex-valued portion; by using the subsequent relation this portion isignored:

$\begin{matrix}{\alpha_{EI} = {~~~}{\arcsin\left( {{{- \lambda}/\left( {\pi \cdot s} \right)} \cdot} \right.}} \\\left. {\arctan\left\lbrack {{Re}\left( {j \cdot {{w\left( {j,l,{n_{Az} + {N/2}}} \right)}/{w\left( {j,l,n_{AZ}} \right)}}} \right)} \right\rbrack} \right) \\{= {\arcsin\left( {{\lambda/\left( {\pi \cdot s} \right)} \cdot} \right.}} \\{\left. {\arctan\left\lbrack {{Im}\left( {{w\left( {j,l,{n_{Az} + {N/2}}} \right)}/{w\left( {j,l,n_{AZ}} \right)}} \right)} \right\rbrack} \right),}\end{matrix}$Re and/or Im representing the real and/or imaginary part of therespective argument.

If in a distance-relative-speed-gate with an azimuth angle there areseveral reflections from different elevation angles (with stronglyexpanded objects and/or because of reflections at the road surface, seebelow), then they cannot be dissolved, i.e. separated by the aboveformula for determining the elevation angle; by a significantcomplex-valued portion in the size j·w(j, l, n_(Az)+N/2)/w(j, l, n_(Az))it can only be recognized that there must be reflections fromsignificant different elevation angles.

The represented approach for the measurement and/or estimation ofelevation angles can be interpreted also in such a way that into thedigital beam formation for the azimuth angle a mono-pulse method for theelevation angle is incorporated (mono-pulse method means that by phasecomparison of two offset antennas (groups) an angle is determined). Thisapproach has the advantage that on the one hand all evaluation methods(such as e.g. the simple digital beam formation with a DFT) based onequidistant receiver antennas can be maintained and that on the otherhand for the azimuth angle there are no losses for the accuracy and onlysmall losses for the separation capability (the latter applies only withobjects with an azimuth angle distance corresponding to half the DFTlength, if for the objects a position outside of the horizontal plane ispossible); with a conventional approach for a simultaneous angularmeasurement in azimuth and elevation, which comprises for the antennacombinations two groups one above the other without horizontal offset toeach other, with an equal number of antenna combinations this wouldresult in that the accuracy and separation capability for the azimuthangle would be halved.

It should be noted that the periodic vertical offset of the antennacombinations can in principle also be embodied with a higher periodlength P than 2. As a result of an object outside of the horizontalplane then in general P power peaks with a respective distance N/P arisein the spectrum, wherein N is the DFT-length of the digital beamformation; from the values of these power peaks again the elevationangle can be determined, wherein now even a separation capability ofobjects via the elevation angle will be possible. By such an approachtwo digital beam formations (for azimuth and elevation) are superimposedin a DFT.

It is to be stressed that the approach presented here for an elevationmeasuring capability is essentially cost-neutral.

In a real environment it is to be considered for the elevationmeasurement that the road surface has a reflecting property; this isshown in FIG. 17. An object receives transmission power on a direct andon a path reflected at the road surface. For the power reflected at theobject and received by the sensor there are equally both ways; thus thesensor in addition to the real object sees a mirror object, which hasapproximately the same (radial) distance as the real object, but lies bythe height h_(o) of the real object underneath the road surface.Dependent on the height h_(s) of the sensor above the road surface thereal and the mirror object have a different amount of the elevationangle, wherein the difference decreases with an increasing distance. Thephase of the received signals of real and mirror object are generallydifferent, since they slightly differ in their distance; this phasedifference changes above the distance r of the real object. Thedescribed effects are the stronger pronounced, the higher the realobject is located above the road surface.

At least for far away objects the real and mirror object are in the samedistance-relative-speed-gate; they have the same azimuth angle, butdifferent elevation angles. For the period length 2 considered above asan example for the vertical offset of the receiver antennas both objectscannot be dissolved; on average the reflection focus lies approximatelyon the height of the road surface. By a significant complex-valuedportion in the proportion V=j·w (j, l, n_(Az)+N/2)/w (j, l, n_(Az)) itcan be recognized, however, that there must be objects in significantlydifferent elevation angles. Since the phase difference between thereceived signals of real and mirror object changes over the distance,over the distance also the complex-valued portion of the proportion Vand thus the size Im(V)/|V| varies, which in the following is to bereferred to as relative imaginary part of V. The distance-relatedvariation of the relative imaginary part of V is the higher, the higherthe real object is located above the road surface. Apart from the closerange this criterion can be utilized for the distinction of relevantobjects on the roadway (e.g. vehicles and pedestrians) and objects whichcan be passed from underneath (i.e. objects above the roadway such ase.g. bridges and signs); e.g. as of a certain distance-related variationof the relative imaginary part of V this points to an object which canbe passed from underneath. FIG. 18 a shows as an example the relativeimaginary part of the proportion V over the distance r for a vehicle(medium height h_(o)=0.5 m, vertical expansion 0.3 m) and a bridge(average height ho=5 m, vertical expansion 0.3 m), wherein the sensorheight is h_(s)=0.5 m.

Furthermore, this criterion can also be used in the closer range todistinguish relevant protruding objects on the roadway (e.g. vehiclesand pedestrians) on the one hand and on the other hand smaller objects(e.g. can of coke) lying on the road and thus being able to be drivenover as well as unevenness of the road surface (e.g. by offset boardjoint). For not or only slightly protruding objects on the roadway thedistance-related variation of the relative imaginary part of V is muchsmaller than for significantly protruding objects. Besides, in the closerange also the actually measured elevation angle can be used, inparticular as by the elevation beam focusing the reflections of a realprotruding object are significantly larger than the reflections of itsmirror object, so that approximately the actual angle of the real objectis measured, from which approximately its real height can be determined.

Apart from the relative imaginary part of the proportion V=j·w(j, l,n_(Az)+N/2)/w(j, l, n_(Az)), in principle also any other measure can beused, which has at least one deviation in relation to the proportionsresulting from only one reflection from an elevation angle, to derivethereof a criterion for recognizing objects which can be passed fromunderneath or above. Thus, e.g. also only the absolute imaginary partIm(V) of the proportion V can be used. The amount of this measure Im(V),in particular filtered via the object distance, and its distance-relatedvariation are the higher, the higher the real object is located abovethe road surface; the filtration of the amount over the object distancecan be linear (e.g. average over a distance section) or nonlinear (e.g.maximum over a distance section). FIG. 18 b shows as an example theimaginary part of the proportion V over the distance r for a vehicle(average height h_(o)=0.5 m, vertical expansion 0.3 m) and a bridge(average height h_(o)=5 m, vertical expansion 0.3 m), wherein the sensorheight is h_(s)=0.5 m.

Furthermore, naturally not only one, but also several measures can beused, which each utilize at least one deviation in relation to theproportions resulting with only one reflection from an elevation angle,in order to derive thereof a combined criterion for recognizing objectswhich can be passed from underneath or above.

The elevation measuring capability can also be used for recognizing andif necessary for correcting a misalignment of the sensor in elevationdirection and/or for monitoring its elevation orientation. Fordetermining the actual elevation orientation only moved objects whichare sufficiently far away are suitable, since moved objects (vehicles)apart from few special cases lie for instance on the same height as theown vehicle and in sufficient distance the road reflections have onlylittle influence on the measured elevation angle, since the elevationangles of the real and mirror object differ only little (how far theobjects have to be away depends on the required accuracy for thedetermination of the elevation orientation. Stationary objects, incontrast, are not suitable since they can lie in different elevationangles (on or above the roadway).

If on average for moved objects far away an elevation angle unequal 0°is measured, then the sensor shows a misalignment by this averagemeasured elevation angle, as other vehicles on average are locatedapproximately in horizontal direction to the own vehicle, i.e. with areal elevation 0°; for example the sensor for an average measuredelevation angle of +2° (for a sensor objects lie approx. 2° above theroadway) looks approx. 2° downward.

The averaging via measured elevation angles of several objects can beeffected either linear, i.e. by weighted averaging, however, nonlinearaveraging is more suitable, which reduce the influence of outliers in aseries of measurements—the median is mentioned as an example.

Outliers in the measured elevation angle with the moved objects used fordetermining the elevation orientation can in principle be avoided to themajority by e.g. the subsequent measures:

-   -   Only objects are taken, which move on the roadway of the own        vehicle; thus no other roadways which are offset in height to        the own roadway can have an influence.    -   The elevation measurement is distorted by reflections at tunnel        slabs and at the bottom sides of bridges; this, however, can be        mostly recognized by a significant complex-valued portion in the        proportion    -    V=j·w(j, l, n_(Az)+N/2)/w(j, l, n_(Az)). Hence such objects are        not to be used. If such objects are recognized, also for safety        reasons for a certain period of time no objects can be used for        averaging.    -    If in the system there are still other tunnel recognition        mechanisms, basically when recognizing a tunnel no object can be        used.

If a misalignment of the sensor in elevation is recognized, this can beeasily out-calibrated; for this purpose only each second of the valuesresulting over the 8 antenna combinations before the third DFT is to beturned by a corresponding phase value.

Embodiments 5-10 According to FIG. 21-26

The embodiments considered so far have only one column per individualantenna (thus per transmitter and receiver antenna), whereby they emitvery wide in horizontal direction (azimuth). Such arrangements aretypically used for close range sensors, since they must have a widehorizontal detection range, but, however, do not have a large reach. Thehorizontal distance d=λ/2 of the receiver antennas to each other isselected so small that the association between the phase differencesΔφ=π·sin(α_(Az)) of adjacent antenna combinations and the azimuth angleα_(Az) in the azimuth range ]−90°, +90°[ is clear (a phase shift causedby the elevation angle with vertically offset antennas, is considerednot here).

Remote range sensors opposite to close range sensors have therequirement of a higher reach and thus a system sensitivity as well as ahigher measurement accuracy and a separation capability for the azimuthangle; in return the horizontal detection a may be restricted. In orderto realize these requirements, the distance of the antennas to eachother is increased (e.g. by the factor 3 in relation to the previousinterpretation, so that the distance between the receiver antennasamounts to d=3λ/2 and the distance between the transmitter antennas to 4d=6λ.) With this on the one hand antennas with several columns and thusstronger focusing in azimuth direction can be realized, resulting in ahigher antenna gain and thus a higher system sensitivity (at the expenseof a reduced azimuthal detection range), and on the other hand theazimuth angle-caused phase differences Δφ=2 πd/λ·sin(α_(Az)) of adjacentantenna combinations have a corresponding greater impact, whichincreases the measurement accuracy and separation capability for theazimuth angle.

The connection Δφ=3π·sin(α_(Az)) resulting for d=3λ/2 between the phasedifference Δφ and azimuth angle α_(Az) is shown in FIG. 19; now a changeof the phase difference of 6π corresponds to the azimuth angle range−90° . . . 90. However, as phases can be measured only up to an unknownintegral multiple of 6π, it comes to ambiguities—thus e.g. it cannot bedistinguished between the azimuth angles −41, 8°, 0° and +41, 8°, asthey are measured with a phase difference Δφ=0.

These ambiguities are avoided, if the individual antennas fortransmission and and/or reception only have an azimuthal detection rangelimited in such a manner that hereto corresponds a change of thedifference phase Δφ of maximum 2π. For the above numerical example thisrequirement can be complied with by the azimuthal detection range −19.5. . . +19.5. In FIG. 20 a a corresponding ideal antenna diagram isshown, which suppresses any transmission and/or reception for azimuthangles outside of this range. In reality such antenna diagrams withsharp detection limits and complete suppression cannot be generatedoutside. FIG. 20 b shows a realizable antenna diagram, which outside ofthe azimuth range −19.5 . . . +19.5 has at least 15 dB suppression. Onsystem level a suppression of the double value 30 dB results, providedthat the antennas have such an antenna diagram both for transmission andfor reception. With this only for very strong reflective objects it cancome to ambiguities for the azimuth angle.

In order to largely avoid the ambiguities for the azimuth angle by anaccordingly limited azimuthal detection range of the antennas fortransmission and/or reception, when seen horizontally the width of theseantennas must be at least about twice as large as the distance d ofhorizontally successive receiver antennas. For compliance with thisrequirement in the following five different approaches for planarantennas are introduced:

-   -   The transmitter antennas when seen vertically are arranged in        another plane (i.e. in another range) than the receiver        antennas, i.e. they lie one above the other (see FIG. 21 as an        example). Thus the horizontal width of the transmitter antennas        can be selected up to four times higher than the distance d of        the horizontally successive receiver antennas. In the example        according to FIG. 21 only the transmitter antennas have the        required narrow azimuthal detection range, however, not the        receiver antennas; thus for stronger reflective objects it can        still come to ambiguities for the azimuth angle. It should be        noted that FIG. 21 represents an example for the fact that form        and emission characteristic of the transmitter antennas may in        principle be different from the form and emission characteristic        of the receiver antennas.    -   The horizontally successive receiver antennas are vertically        offset in such an alternating manner that they lie in E        different vertical planes; in the illustrated example according        to FIG. 22 there are two planes, the alternating offset thus has        the period length 2—the two receiver antennas RX0 and RX2 lie        one plane above the other two receiver antennas RX1 and RX3.        When using E different planes for the receiver antennas their        horizontal width can be selected up to E-times larger than the        distance d of horizontally successive receiver antennas. As a        result of the vertical offset of the receiver antennas        implicitly the above described elevation measuring capability        ensues.    -   Transmitter and/or receiver antennas are not arranged        vertically, but inclined to the horizontal (see FIG. 23 as an        example, where one of the inclined antennas is used for        transmission and reception). Thus the horizontal width of        antennas can be made nearly arbitrary larger than its distance.        However, the high focusing of the emission characteristic is        only for elevation angles within the range of 0°, i.e. for        objects approximately in the horizontal plane—however, apart        from the absolute close range at this place lie all objects        relevant from the system point of view. For elevation angles        significantly deviating from 0° there is less beam focusing in        the horizontal direction, which can have a negative effect at        best in the absolute close range and for objects located across        the road. The smaller beam focusing for elevation angles        deviating significantly from 0° can principally be avoided on        system level by the fact that transmitter and receiver antennas        are tilted in different directions (the ones to the right, the        others to the left).    -   Antennas are locked into each other in horizontal direction (see        as examples FIGS. 24 a and 24 b, where the receiver antennas are        locked into each other). Thus the horizontal width of antennas        can be made twice as large as their distance. The locking in the        case of the arrangement according to FIG. 24 a results in an        offset of the antennas in vertical direction and thus implicitly        in the above described elevation measuring capability, with the        alternative antenna assembly according to FIG. 24 b this is not        the case. Instead of the locking shown in both examples from the        side, it could be embodied also from above and/or from        underneath.    -   Emitting and/or receiving elements are jointly used at least by        two antennas (see FIG. 25 as an example, where by two adjacent        receiver antennas each three patch columns are jointly used and        their received power is distributed to the two antennas).        Theoretically, antennas can be made thereby nearly arbitrary        wider than for their distance, however, in practice in this case        the power distribution networks get more and more complicated,        in particular because crossings of high frequency connections        are difficult to be realized.

Without the approaches for the arrangement of the antennas, representedabove, only a horizontal width of the antennas could be realized, whichcorresponds to its distance which would result in strongly pronouncedambiguities for the azimuth angle.

It should still be mentioned that the above antenna assemblies forremote range sensors always comprise two transmitter antennas on theoutside and thereby the effective aperture of the sensor can be nearlydoubled in relation to its width—by means of this also in the 24GHz-range sensors for long-reach functions with an acceptable sensorsize can be realized. In order to be able to receive received signalsfrom objects up to a distance of 200 m, the linear frequency modulationin contrast to the interpretation for the embodiments 1 and 2 have onlyhalf the frequency deviation, i.e. 93.75 MHz.

The above described methods for realizing an overlapping of antennas, inparticular in horizontal direction, can be applied also for close rangesensors. The embodiment 2 according to FIG. 10 is an example for anarrangement of the transmitter antennas above the receiver antennas(i.e. when seen vertically they are in different planes). In FIG. 26 ahorizontal locking of the transmitter antennas with the receiverantennas is shown, by which a distance of d/2=λ/4 can be realizedbetween the transmitter antennas and the respectively adjacent receiverantennas, whereby despite an arrangement of the transmitter and receiverantennas essentially in one plane no common transmitter and receiverantenna as in the arrangement 3 according to FIG. 12 is required.

Embodiment 11 According to FIG. 27

The measures represented above for reducing and/or avoiding ambiguitiesfor the azimuth angular measurement with remote range sensors have ledto a reduced azimuthal detection range. If, however, simultaneouslylong-reach and short-reach functions are to be realized now with asensor, then a wide azimuthal detection range can be required (forexample −80° . . . +80). This can be realized by the subsequentlyrepresented approach, which is explained on the basis of the embodiment11 according to FIG. 27.

In contrast to the antenna configuration according to FIG. 23 with theconfiguration according to FIG. 27 considered now there are twodifferences: on the one hand the individual antennas significantly emitand/or receive also outside of the azimuthal range −19.5 . . . +19.5 bycorresponding shoulders in the antenna diagram (see FIG. 28); suchshoulders can be generated by the fact that patches in the centralportion when seen horizontally of the individual antennas emitsignificantly more than the others, whereby a narrow and a wide antennadiagram are superimposed (by such an approach the antenna gain and thusthe system sensitivity are reduced in the central portion, where a highreach is required, only relatively small). And on the other hand the tworeceiver antennas RX0 and RX2 are offset to the right by the horizontaldistance t=λ/8, which is smaller by the factor 12 than the centralhorizontal offset d=3λ/2 of the receiver antennas. Thus the relativephase centers of the 8 antenna combinations do not lie equidistantly inhorizontal direction, but they have an offset alternating with theperiod length 2 to an equidistant raster. For a far away object restingrelative to the sensor with an azimuth angle α_(Az) therefore the phaseφ(m) of the received signals has a portion alternating with the periodlength 2 in addition to a linear portion over the antenna combinationsm=0, 1, . . . 7 and analogue to FIG. 7 b this results inφ(m)=φ(0)+2π/λ·sin(α_(Az))·(d·m−t·mod(m,2)).

Provided that in the assigned distance-relative-speed-gate (j, l) thereis only this object, the there complex values v(j, l, m) after thesecond DFT over the 8 antenna combinations m=0, 1, . . . 7 result inv(j,l,m)=K·exp[j·(φ(0)]·exp[j·2π/λ·sin(α_(Az))·d·m]·exp[−j·2π/λ·sin(α_(Az))·t·mod(m,2)],wherein K is the constant amount of these values.

Thus one receives analogue proportions as with the above consideredelevation measuring capability by a vertical offset alternating with theperiod length 2 of the individual receiver antennas. By the factorexp[−j·2π/λ·sin(α_(Az))·t·mod(m, 2)], which is generated by thehorizontal offset alternating with the period length 2 of the individualreceiver antennas, the spectrum w(j, l, n) of v(j, l, m) formed in thethird DFT has two power peaks—one power peak at the frequency n_(Az)corresponding to the azimuth angle α_(Az) and one at the frequencyn_(Az)+N/2 distanced by half the DFT-length, wherein for the proportionof the spectral values of these two power peaks it applies:w(j,l,n _(Az))/w(j,l,n _(Az)+N/2)=(1+exp[−j·2π/λ·sin(α_(Az))·t])/(1−exp[−j·2π/λ·sin(α_(Az))·t]);

FIG. 29 shows an exemplary course of the amount of the spectrum w(j, l,n) with two power peaks distanced by half the DFT-length N=16 with n=2and n=10.

For all azimuth angles α_(Az) in the detection range −90° . . . +90°with the here considered t=λ/8 the proportion w(j, l, n_(Az))/w(j, l,n_(Az)+N/2) with regard to the amount is larger than one. Thus thefrequency value n=n_(Az) belonging to the azimuth angle α_(Az) can bedetermined as the position of the power peak being higher with regard toits amount; in the example according to FIG. 29 thus n_(Az)=2 results.However, with this the azimuth angle α_(Az) is not yet clearly defined,as to each n=n_(Az) three different values of α_(Az) correspond, thismeans in the example according to FIG. 29 still the three azimuth anglesα_(Az)=4.7°, 48.6° and −35.7° come into question. By transformation ofthe above relation (analogue to the above derivation for the elevationmeasuring capability), however, now the azimuth angle can be alsodetermined from the proportion w(j, l, n_(Az))/w(j, l, n_(Az)+N/2):α_(Az)=arcsin(−λ/(π·t)·arctan [j·w(j,l,n _(Az) +N/2)/w(j,l,n _(Az))]).

As this relation for t=λ/8 is clear over the entire azimuthal detectionrange −90° . . . +90° (because of 2π/λ·sin(α_(Az))·t ε ]−π/4, π/4[),this allows for determining the right one of the three azimuth anglescoming into question—the azimuth angle determination is thus clear for asingle object in a distance-relative-speed-gate, wherein the ambiguityis generated via the alternating horizontal offset of the receiverantennas (without this offset it could not be distinguished in each casebetween three various azimuth angles).

If in a distance-relative-speed-gate there are reflections of severalobjects in different azimuth angles, which correspond to two frequencyvalues n and n+N/2, then they generally can no longer be separated; by athen generally significant complex-valued portion in the size j·w(j, l,n_(Az)+N/2)/w(j, l, n_(Az)) it can only be recognized that there areseveral of such objects and certain possible hypotheses can be derived.Which of this hypothesis is the right one, can be mostly recognized byplausibility considerations; examples for this are:

-   -   With the aid of the tracking (i.e. the observation of objects        over several cycles) with objects moved relative to the sensor        it can be recognized in general, which one of several possible        hypothesis is the right one (e.g. only temporary a fusion on the        same frequency values n and n+N/2 takes place and mostly only        for one hypothesis a useful course of the object results).    -   By an antenna diagram as is represented in FIG. 28 of the        individual antennas objects from the central portion with a high        antenna gain generally have significantly more level than        objects from the outer portion with a smaller antenna gain        (shoulders), so that with the presence of an object in the        middle and one in the outer portion the middle object dominates        in the level and is therefore recognized with the proper azimuth        angle, which is sufficient from functional view, since this        middle object is then the relevant one.

The example according to FIG. 27 considered so far did not have avertical offset in the receiver antennas and thus no elevation measuringcapability. In principle also such a vertical offset can naturally besuperimposed. If different period lengths are selected for the verticaland the horizontal offset, then their effects can be separated directly,so that azimuth and elevation angle for a single object in adistance-relative-speed-gate can be clearly determined; with the sameperiod length corresponding plausibility considerations are necessary.

The explained approach for the clear measurement of the azimuth anglewith widely emitting remote range sensors can be interpreted also insuch a way that in the high-resolution and accurate, but ambiguousdigital azimuth beam formation a mono-pulse method is incorporated foran azimuth angle determination which for a single object is clear butrough. This approach has the advantage that on the one hand allevaluation methods based on equidistant receiver antennas (such as e.g.the simple digital beam formation with a DFT) can be maintained and thaton the other hand for the azimuth angle there are no losses for theaccuracy and only small losses for the separation capability (the latteronly with objects, which generate power peaks with identical frequencyvalues n). In order to be able to determine the azimuth angle asaccurate as possible from the higher of the two power peaks, aninterpolation with adjacent power values can be performed and from thisthe interpolated maximum can be determined—as is already mentionedabove.

It should be noted that the periodic horizontal offset of the antennacombinations can principally be also embodied with a higher periodlength Q than 2. By an object then generally Q power peaks withrespective distance N/Q result in the spectrum, wherein N is theDFT-length of the digital beam formation; from the values of these powerpeaks again the azimuth angle can be clearly determined, wherein noweven a separation capability of objects, which generate power peaks withidentical frequency values n, will be possible. By such an approach twodigital beam formations in a DFT are superimposed—the one is fine, butambiguous, the other for single objects is clear, but rough.

It is to be stressed that the approach introduced here for a sensor forrealizing short- and long-reach functions, is essentially cost-neutral.

Final Remark

It should be noted that the considerations and embodiments according tothe invention represented on the basis of the above examples can betransferred to general calculations and parameter interpretations, i.e.they can be applied also for other numerical values (e.g. for thehorizontal and vertical distance of the antennas). Therefore, often alsogeneral parameters are indicated in formulas and images apart from theconcrete numerical values.

The invention claimed is:
 1. A radar system for recording theenvironment of a motor vehicle, comprising: a number N_(S) oftransmitter antennas adapted to emit transmission signals; a numberN_(E) of receiver antennas adapted to receive the transmission signalsreflected from objects in the environment; and signal processingcircuitry connected and adapted for processing the received signals;characterized in that the antennas are arranged so that a phase centerof at least one receiver antenna does not lie, with regard to a spatialdirection R, outside phase centers of two transmitter antennas that areoffset in said spatial direction, the signals received by said receiverantenna are separated according to the two portions originating fromsaid two transmitter antennas, received signals from differentcombinations of the N_(S) transmitter antennas and of the N_(E) receiverantennas are acquired, wherein for each combination a relative phasecenter is defined as a sum of the two vectors from a reference point tothe phase centers of the respective transmitter antenna and of therespective receiver antenna and the relative phase centers of thesecombinations of transmitter and receiver antennas lie at leastapproximately equidistant with regard to the spatial direction R, withregard to the spatial direction S which runs perpendicular to thespatial direction R, the position of the relative phase centers of thesecombinations of transmitter antennas and receiver antennas variesperiodically with the period length P, if a sequence of thesecombinations of transmitter antennas and receiver antennas isconsidered, which is arranged in the spatial direction R with regard tothe position of the relative phase centers, and in the signal processingcircuitry it is utilized that the received signals of an objectdependent on its angular position in the spatial direction S have aphase portion alternating periodically with the period length P over thecombinations of transmitter and receiver antennas arranged in suchmanner, whereby for this spatial direction S assertions regarding theangular position of objects and/or regarding a misalignment of the radarsystem are possible.
 2. A radar system according to claim 1, in whichthe spatial direction R lies horizontally and the spatial direction Slies vertically, and in the signal processing means, for recognizing inparticular stationary objects which can be driven over or under, ameasure is used, which utilizes at least one deviation in relation tothe proportions resulting with received signals from only one elevationangle, wherein in connection with the reflecting property of roadsurfaces the size and/or the amount, in particular filtered in each caseover the object distance, and/or a distance-related variation of thismeasure is used for an at least rough estimate of the height of objectsabove the road surface.
 3. A radar system according to claim 1, whereinthe signal processing circuitry is adapted to make said assertionsregarding the misalignment of the radar system.
 4. A radar system forrecording the environment of a motor vehicle, comprising: at least twotransmitter antennas adapted for emitting transmission signals; one ormore receiver antennas adapted for receiving the transmission signalsreflected by objects in the environment; and signal processing circuitryconnected and adapted for processing the received signals; characterizedin that the antennas are arranged so that a phase center of at least onereceiver antenna does not lie, with regard to a spatial direction R,outside phase centers of two transmitter antennas that are offset insaid spatial direction, the signals received by said receiver antennaare separated according to two portions originating from said twotransmitter antennas received signals from different combinations oftransmitter and receiver antennas are acquired, wherein for eachcombination a relative phase center is defined as a sum of the twovectors from a reference point to the phase centers of the respectivetransmitter antenna and of the respective receiver antenna, transmitterand receiver antennas used thereby each have at least approximately thesame emission characteristic, wherein the emission characteristic ofthese transmitter antennas can be different from the emissioncharacteristic of these receiver antennas, with regard to the spatialdirection R the position of the relative phase centers of thesecombinations of transmitter and receiver antennas varies periodicallywith the period length Q by an equidistant raster, and in the signalprocessing circuitry, for determining the position of objects in thespatial direction R, it is utilized that the received signals of anobject dependent on its angular position in the spatial direction Rapart from a linear phase portion have a phase portion alternating withthe period length Q, if a sequence of the combinations of transmitterand receiver antennas ordered in the spatial direction R with regard tothe position of the relative phase centers is considered.
 5. A radarsystem for recording the environment of a motor vehicle, comprising: aplural number N_(S) of transmitter antennas that are adapted to emittransmission signals; a plural number N_(E) of receiver antennas thatare adapted to receive the transmission signals reflected from objectsin the environment, and that are arranged with phase centers thereofspaced apart from one another by a spacing distance d in a spatialdirection R; and signal processing circuitry connected and adapted forprocessing the received signals, characterized in that said antennas arearranged so that a phase center of at least one said receiver antennadoes not lie, with regard to said spatial direction R, outside of phasecenters of two said transmitter antennas that are offset from oneanother in said spatial direction R, the signals received by said onereceiver antenna are separated according to two signal portionsoriginating from said two transmitter antennas, the transmitter antennasare arranged with phase centers thereof spaced apart from one another insaid spatial direction R by a spacing distance (Kd) equal to an integralfactor K times the spacing distance d of the receiver antennas wherein Kmust be less than or equal to the number N_(E) of the receiver antennas(K≦N_(E)), the transmitter antennas each have at least approximately thesame emission characteristic, the radar system synthesizes a virtualarrangement with one transmitter antenna and a maximum of N_(S)×N_(E)receiver antennas that are equidistantly arranged in the spatialdirection R and that each have an at least approximately identicalemission characteristic, the N_(S) transmitter antennas and the N_(E)receiver antennas are realized in planar technology and are arranged ona plane surface, and at least two of the N_(S) transmitter antennas andof the N_(E) receiver antennas overlap with regard to the spatialdirection R, wherein this overlap is realized by at least one of thefollowing features of the N_(S) transmitter antennas and the N_(E)receiver antennas: a) the antennas are offset to each other with regardto the spatial direction S which is perpendicular to the spatialdirection R, b) the transmitter and/or the receiver antennas have aninclined form with regard to the spatial direction R, c) the antennasare intermeshed with each other with regard to the spatial direction R,d) emitting and/or receiving elements are jointly used by at least twotransmitter and/or receiver antennas, and/or e) at least one antenna isused both for transmission and reception.
 6. A radar system according toclaim 5, in which, with regard to the spatial direction R, the width ofthe N_(S) transmitter antennas and/or the width of the N_(E) receiverantennas is larger than the distance of the N_(E) receiver antennas toeach other.
 7. A radar system according to claim 5, wherein the overlapis realized by at least said feature a).
 8. A radar system according toclaim 5, wherein the overlap is realized by at least said feature b). 9.A radar system according to claim 5, wherein the overlap is realized byat least said feature c).
 10. A radar system according to claim 5,wherein the overlap is realized by at least said feature d).
 11. A radarsystem according to claim 5, wherein the overlap is realized by at leastsaid feature e).
 12. A radar system for recording the environment of amotor vehicle, comprising: a plural number N_(S) of transmitter antennasthat are adapted to emit transmission signals, and that each have atleast approximately the same emission characteristic; a plural numberN_(E) of receiver antennas that are adapted to receive the transmissionsignals reflected from objects in the environment, that each have atleast approximately the same reception characteristic, and that arearranged with phase centers thereof spaced apart from one another by aspacing distance d in a spatial direction R; and signal processingcircuitry connected and adapted for processing the received signals,characterized in that said antennas are arranged so that a phase centerof at least one said receiver antenna does not lie, with regard to saidspatial direction R, outside of phase centers of two said transmitterantennas that are offset from one another in said spatial direction R,the signals received by said one receiver antenna are separated into twosignal portions respectively originating from said two transmitterantennas that are simultaneously emitting their respective transmissionsignals with respective phase modulations that are different from oneanother, wherein said two signal portions received by said one receiverantenna are separated from one another based on said different phasemodulations thereof, and the transmitter antennas are arranged withphase centers thereof spaced apart from one another in said spatialdirection R by a spacing distance (Kd) equal to an integral factor Ktimes the spacing distance d of the receiver antennas wherein K must beless than or equal to the number N_(E) of the receiver antennas(K≦N_(E)), whereby the radar system synthesizes a virtual arrangementwith one transmitter antenna and a maximum of N_(S)×N_(E) receiverantennas that are equidistantly arranged in this spatial direction R andthat each have an at least approximately identical emissioncharacteristic.
 13. A radar system according to claim 12, wherein in thesignal processing circuitry the position of objects in the spatialdirection R is determined by the fact that for the angle formation withregard to this spatial direction R a digital beam formation orhigh-resolution methods are used.
 14. A radar system according to claim12, wherein one of said transmitter antennas is also connected to saidsignal processing circuitry and is operative for receiving saidtransmission signals reflected by said objects, whereby a singlephysical antenna is both one of said transmitter antennas and one ofsaid receiver antennas.
 15. A radar system according to claim 14,wherein all of said transmitter antennas and said receiver antennas arearranged with said phase centers thereof aligned with each other alongsaid spatial direction R.
 16. A radar system according to claim 12,further comprising a signal generator adapted to generate saidtransmission signals, wherein said plural number N_(S) consists of twoof said transmitter antennas that are connected to said signal generatorfor emitting the transmission signals and that are not connected to saidsignal processing circuitry and are not operative for receiving saidtransmission signals reflected by said objects, wherein said receiverantennas are arranged with said phase centers thereof aligned with eachother along a first imaginary line in said spatial direction R, andwherein said transmitter antennas are arranged with said phase centersthereof aligned with each other along a second imaginary line in saidspatial direction R and offset from said phase centers of said receiverantennas in a spatial direction S perpendicular to said spatialdirection R.
 17. A radar system according to claim 12, wherein each oneof said antennas comprises a plurality of patch antenna elements thatare connected to one another in at least one linear row of said patchantenna elements extending along an imaginary line having a directionalcomponent perpendicular to said spatial direction R.
 18. A radar systemaccording to claim 17, wherein each one of said antennas consists of asingle linear row of said plural patch antenna elements.
 19. A radarsystem according to claim 17, wherein each one of said antennascomprises an interconnected plurality of said linear rows of said pluralpatch antenna elements.